Monday, June 13, 2011

RF Audio Link

CHAPTER 1: INTRODUCTION

1.1 MOTIVATION
Wireless headphone is a great convenience item for the remote audience to stay connected to the selected audio source and still listen to music on headphones while moving or not physically attached to the audio source. The commonly used medium of wireless transmission methods is either Infra-red or Radio Frequency.

“Stereo Infra-red headphones” for household AV system have been fully operational in domestic markets for many years but has a number of undesirable drawbacks, such as:

• Short distance of transmission.
• Narrow coverage angle and area.
• Transmission and reception must be in direct line of sight, i.e. limits the freedom of movement.
• Interference and hiss will be heard in the existence of obstacles.
• Cannot generally be used in direct sunlight and fluorescent lighting, which produces infra-red noise.

However, wireless headphone through RF (Radio Frequency) out-performs the commercially available Infra-red headphone by increasing the range, widening the coverage area and particularly, direct line of sight transmission is not relevant, i.e. enabling users to move freely between partitions and rooms or even out-doors.

Low cost RF headphones for consumer applications have predominantly used the lower VHF frequency bands for unlicensed low-power RF operation. These frequencies, such as 36.4Mhz, 37.1Mhz, 40.68Mhz and 49.9Mhz, have been used by some manufacturers for the reasons of large supply of low-cost highly integrated circuits. Furthermore, the over-crowded conditions that exist in the traditional VHF frequency bands and the need for additional bandwidth to support high fidelity and advanced products using digital technology have motivated electronic circuit designers to develop products and explore the feasibility that can operate in the spectrum of UHF region, such as the European 433/869 MHz ISM bands or the U.S. 902-928 MHz ISM band.

Due to the fast growing and exponential development of highly integrated monolithic RF transceiver products, microchips and SMT manufacturing technology, the recent trend in the RF and microwave industry has been a move towards increasing the number of components realized on one radio frequency integrated circuit. This extensively reduces the number of components used and achieves a more compact and reliable RF products, such as “RF Stereo Headphone”. However, this trend has resulted in complex circuit integration of which often requires reactive as well as other circuit components to be supplied in the form of an external circuit. Because the manufacturer’s suggested circuit is often developed with a specific application in mind, the same circuit may not satisfy the demands of another application. This will hinder and constrain the innovation of a new design.

In 1994, Dr. K.F. Tsang, Managing Director of CityCom Technology Ltd; a subsidiary of CityU Enterprises Ltd, who contributed to the research and design of a 900Mhz wireless stereo headphone project. By using the SMT technology, it not even improved the product yield but achieved superior sound quality as well. Nowadays, many local and foreign manufacturers have their own technology and design of wireless stereo headphones in different RF bands. The following is a typical specifications and outlook of a model RS860 manufactured by a local manufacturer “Uni-Art Precise Products Ltd”.

















To make the products as compact and low cost as possible, all these products are shifted and operated in UHF region like 434Mhz, 863Mhz, 900Mhz or even the 2.4GHz. Many efforts are underway to increase the integration level of the transceiver. The ultimate goal would be a single chip RF transceiver in a single technology with a minimum number of off-chip components, that is, an antenna to receive or transmit the RF signal, a power supply, and a crystal reference to provide a clean frequency reference. Another important single chip stereo signal encoder, either by analogue or digital method can then be directly interfaced with the single chip RF transceiver, such as BA1404 by RHOM. With high integration level, this will largely reduce the total components count, especially, achieve lower cost, and lower power dissipation. Furthermore, for these stereo headphone products, traditional analogue FM technology is still being used within the single chip RF transceiver and digital method (time division multiplexing technique) is the most efficient way to be implemented in the single chip stereo signal encoder, but at the expense of low channel separation.

Despite the great benefits of compact in size, lower cost, lower power dissipation by using the present high-end technology. In view of the specifications and outcomes of the existing products, several drawbacks and limitations can be observed as follows:

• Hazardous radiation to human body in UHF or microwave region is unknown but still be a major issue.
• Working in microwave region, means transmission is direct line of sight and suffers significant attenuation in the presence of obstacles.
• Transmitter and receiver are sold in pair, separate receiver may not be sold individually, i.e. High equipment cost for multiple audience.
• For a same model, max 2-3 selectable transmission carriers in the same band are not enough to avoid interference and congestion.
• Remote selection of different audio sources is commonly not available.
• Limited use on dedicated headphones only, not fully compatible with existing Hi-Fi System.
• Low channel separation (Typical 30dB).

The drawbacks of the existing UHF headphones reveal the major inherent issues of compatibility, portability, control and channel separation. However, the shortcoming provides further headroom for this project to explore and investigate. The work presented here is focused on investigating, designing strategies to implement and construct a RF Stereo Audio Link by using suitable modulation principles for all household AV equipment. Nevertheless, which enhances the features and the shorting-comings existed in the available RF headphones that are described above.

1.2 PROJECT OBJECTIVES
In order to overcome the constraints and bearers of the marketed RF headphones as briefed in Section 1.1, the purpose of this project is to develop a universal RF audio link. Under the control of a hand-held remote control unit, different domestic AV sources can be selected. By using appropriate modulation techniques that the selected audio signal can be transmitted through the RF link. Recovery of the original signal can be achieved by a suitable receiver, and the sound will be finally presented on a headphone or through the other remote AV system. The main advantages that this RF audio link are enabling users to move freely between partitions and rooms while listening to the audio/Hi-Fi sources such as radios, CDs or TVs etc, through which outstanding stereo sound still be achieved.

Under those primitive criterions mentioned above, the final product should have the following features:

• Reception through ceilings and walls.
• Reception range is more than 300 meters in open space.
• Provide 4 inputs RF remote control that can select different audio sources, such as CD players and TV-sets, etc.
• The audio link must be in stereo.
• Left/Right channel separation should be better than 30dB.
• Move freely around in-house or outdoors.
• Minimum audio bandwidth should comply with the broadcast standard, i.e. 20Hz to 15KHz.
• RF link ensures excellent sound reproduction

1.3 CHOICE OF OPERATING FREQUENCIES
In Hong Kong, according to the regulations setout by Office of the Telecommunications Authority (OFTA), no wide-band RF audio products can be allowed to operate in any frequency bands.

In Europe, adopting the Recommendations from the European Telecommunications Standards Institute (ETSI), ERC RECOMMENDATION 70 03 at March 2001 sets out the general position on common spectrum allocations for Short Range Devices (SRDs) for countries within the CEPT. Frequency band 863-865MHz is reserved for the applications for wireless audio systems include the cordless loudspeakers; cordless headphones and portable devices, for example portable CD, cassette or radio devices carried on a person.
In the US, users can operate without a license in the license free band such as 900MHz and 2.4 – 2.5GHz so long as they use equipment that has been approved by the FCC. Whereas, unlicensed operation on the FM broadcast band 88-108Mhz is permitted for some extremely low powered devices covered under Part-15 Sec. 15.239 of the FCC's rules.

The regulations for unlicensed operation on the FM broadcast band include:

• The 200 kHz band shall lie wholly within the frequency range of 88-108 MHz.
• The field strength of any emissions shall not exceed 250uV/m at 3m.
• The field strength of any emissions radiated on any frequency outside of the specified 200 kHz band shall not exceed the general radiated emission limits in Sec. 15.209.
• A custom built telemetry intentional radiator operating in the frequency band 88-108 MHz and used for experimentation by an educational institute need not be certified provided the device complies with the standards in this part.

There are much more in the Part 15 Rules. FM broadcast band usage is specifically addressed in Rule No. 15.239.

The most important issue that has to be investigated is the selection of a feasible RF carrier frequency that the Control Link and the Audio Link will use. The performance of these RF Links is very dependent on the RF carrier frequency; it’s radiated power, the actual usage and environment of the system. However, in order to minimize the interference to/from other systems operating in the same or adjacent bands and whether it will conflict with the rules and laws of local authorities, much attention should be paid on the selection of the right operating frequency and the maximum allowable RF radiation power.

As we know, most of the latest RF stereo headphones are designed and operating in the UHF band for achieving a wider bandwidth and high quality of sound. However, standard VHF broadcast transmission standard using the 88MHz to 108MHz still has it’s advantages for providing high-quality sound reproduction, and vast of high quality domestic and portable receivers available in the market. Hence, the final choice for the Audio Link of this project is in the VHF broadcasting band of 88MHz to 108MHz, the justifications are:

• System fabrication in VHF range is easy to handle than that in UHF region, i.e. difficult SMT process and techniques can be eliminated.
• Although unlicensed transmission using the board-casting band in Hong Kong is strictly prohibited. However, by suitable design of RF radiated power and antenna orientation, and used in a confined space, can keep the interference to great extent for the purpose of project and experiment. For the issue of legal and long-term operation of this product, I suggest it should only be used in some legal and unlicensed countries, such as USA or UK.
• Low construction and operational costs, since concerned components can be sourced in the local market place.
• High portability and compatibility with the commercially available portable and desktop FM receivers, i.e. save cost and effort to built receiver in limited time span.
• Available 200KHz bandwidth for each channels, retains high fidelity of sound and music.
• 100 available channels within the 88MHz to 108MHz frequency band, i.e. higher probability of using idle channels within a particular location and environment to avoid interference and congestion.
• Varieties of high quality FM receiver can be available in market place; users can have their own choice to match their own need and preference.

For the Control Link, it operates in 433.92MHz, which is license-exempted for ‘short range device’ such as security control, telemetry and alarm when used in Europe. Refer to The European Telecommunications Standards Institute’s (ETSI) ERC RECOMMENDATION 70 03 (2001), RELATING TO THE USE OF SHORT RANGE DEVICES (SRD) for countries within the CEPT.

The justifications that I decide to use this frequency as the Control Link are:

• Varieties of small-sized 433.92MHz AM or FM Transmitter/Receiver pre-built modules are available.
• Facilitate a compact and simple circuit design.
• Interfacing value-added control facilities is easy to implement.
• Low cost.

1.4 CHOICE OF MODULATION METHODS
In order to convey information through the RF links, low frequency audio or control signals must be impressed on to the RF ‘carrier’ and this process is called modulation. Different types of modulation methods has it’s own characteristics, features, pros and cons. Choice of modulation methods may influence the ultimate circuit design, the detection and recovery algorithm, the efficiency of radiated power, the signal bandwidth, noise and more importantly, the system performance in particular applications, see William Schweber’s [1999] Electronic Communication System.

In my project, Frequency Modulation (FM) is proposed to implement in the Audio Link as broadcast grade of transmission. The reasons for the selection of the FM lie in their efficient use of transmitted power, low power modulation, and excellent noise immunity, by which the 200KHz bandwidth achieves the high quality transmission of audio and music programme. For a full discussion of the signal processing issues and an analysis of the suitability of FM radio signals for use in broadcast transmission, see Gary M. Miller’s [1996] Modern Electronic Communication and William Schweber [1999] Electronic Communication Systems.

For the Control Link, Amplitude Modulation (AM) is proposed for its simplicity and ease of signal recovery. The major drawback by using AM is that the AM signal is prone to noise and interference, hence degrades the quality of the control signal, see Gary M. Miller’s [1996] Modern Electronic Communication and William Schweber [1999] Electronic Communication Systems. Whereas, using an appropriate line coding method such as bi-phase Manchester coding scheme, and data-decoding scheme after demodulation, such as data validation by checking and comparing the received data for a specific number of times to minimize errors, which I will describe it later.


CHAPTER 2: BASIS OF RF COMMUNICATIONS SYSTEM

2.1 BUILDING BLOCK RF COMMUNICATIONS SYSTEM
In broader sense, electronic communication refers to the ways of sending, receiving and processing of information, such as speech, music, and video or computer data by electrical means. The medium through which the information can be conveyed is either a physical wire, or free space. However, RF communications system is focused in using radio frequency as a means to transmit information from one place to another remote place through the medium of space.

In order to be familiar with the RF systems, it is first necessary to have a basic understanding of the configuration and architecture of such systems, background principles and theories of modulations, oscillators and amplifiers, and the building blocks of all related electronic processing circuits and equipment.

Generally, RF systems may be depicted by the following diagram shown in Figure 2.1. Regardless of what the applications, for this project, it concentrates in the particular applications of RF Links for conveying the audio and digital control information. However, all the RF link systems generally involve three major subsystems, they are the transmitter, the transmission medium, and the receiver. The information from the source, which can be either analogue or digital form, depending on the particular system design and requirements, such as audio or control data, is represented by the information input signal m(t). The information finally delivered to the terminal user is denoted by m’(t), which may not be the same as the transmitted information m(t). This may be due to noise n(t) injected in the transmission medium, or some impairments through out each part of the system, such as undesired filtering, non-linearities of electronic components, natural electrical disturbance, e.g. lightning, or man-made noise, such as starting of motor or nearby switching equipment.





The signal-processing block in the transmitter conditions the source signal for more efficient and effective transmission. For example, in an analogue system, it involves the band-pass filter to restrict the bandwidth of m(t); signal level conditioning and impedance transformation for perfectly matching with the subsequent modules. In multiplexed system, which may involve frequency-division-multiplexer (FDM) to allocation different information source in particular frequency spectrum. In a hybrid system, the signal processing may involve an analogue-to-digital converter (ADC), which produces a “digital code word” that represents samples of the analogue input signal. In this project, the information input signals are encoded audio signal containing the stereo information or base-band digital control data.
The processed signal usually contains low frequency information content, which will be unsuitable for immediate transmission. The transmitter carrier circuit then converts the processed signal into a distinct frequency band, i.e. the processed signal is impressed on a higher frequency carrier sine wave, S(t) which is appropriate for the transmission medium of the channel, this process is called modulation.

With regard to this case, the communication channel is air, which will be inevitably introduce some constraints during the wave propagation, such as obstacles blocking and attenuating the transmitted signal, multi-paths between transmitter and receiver that have different time delays and attenuation characteristics, which causes signal fading at the channel output.

The receiver gets the corrupted signal at the channel output and converts it to a base-band signal through the carrier circuits, this process is called demodulation; that can be handled by the subsequent signal processing stages, which amplify and condition the demodulated signal and finally deliver the source information m’(t) to the terminal user.

2.2 USABLE RADIO FREQUENCY SPECTRUM
Radio signals are kind of electromagnetic waves, which propagation characteristics are not only dependent on their frequency and properties but also governed by the environment, such as ambient temperature, atmospheric pressure, humidity and levels of air ionization etc. In the present of the earth and atmosphere, different frequencies would have different propagation characteristics.


RF communications systems are often categorized by the frequency of the carrier. The usable spectrum of radio frequencies extends from the range of 30Hz and increasing to the range of 300GHz. Table 2-1 tabulates the designations and typical use for various frequency bands in the radio frequency spectrum.






CHAPTER 3: BACKGROUND THEORY OF MODULATION

3.1 INTRODUCTION
The primary or sole purpose of most RF communication systems is to transfer or convey information from one location to another through the space. In section 2, I have briefly described the components of a typical RF communication system. One of the very important components is the Modulator, which acts to modulate the information signal, producing at its output the modulated waveform. Modulation is the process hereby, means to systematically use the information signal to vary some parameters of the higher-frequency carrier signal. The carrier sine wave can be represented by vc = Vc sin (2 π fc t + θ) where the parameters are defined as below:

vc: The instantaneous voltage of the carrier signal
Vc: The amplitude of the carrier signal
fc: The frequency of carrier signal
θ: The phase of the signal

It is obvious that there are three different parameters of a carrier can be modified so as to allow for carrying information; the amplitude, frequency or phase of the carrier can be altered systematically by the information signal. Therefore, the type of modulation may be categorized as amplitude modulation (AM), frequency modulation (FM) or phase modulation (PM) respectively. Both AM and FM will be implemented in this project, so a brief understanding of the characteristics for the amplitude modulation and frequency modulation is the goal of this chapter.

Generally speaking, there are two reasons for the need of modulation, they are:

• Concerns of interference -- Direct transmission of information signals would result in great interference problems, since the radio waves would be in the same frequency spectrum, such as voice or music in the band of 20Hz to 15KHz.
• Concerns of antenna efficiency -- Efficient transmission and reception of most information signals at low frequencies is not practical due to large antenna size. To reduce the wavelength for efficient transmission and reception (the optimum antenna size is ¼ of a wavelength). A typical audio frequency of 4000 Hz will have a wavelength of 75 km and would need an effective antenna length of 18.75 km! By comparison, a typical carrier for FM broadcasting is 100 MHz, with a wavelength of 3 m, and could use an antenna only 80 cm long.

3.2 AMPLITUDE MODULATION
Amplitude modulation is the simplest type of modulation method. The information signal, vm is used to vary the amplitude of the carrier, vc to produce a modulated signal, vAM. Here are the three signals in mathematical form:
• Information signal: vm =Vm sin (2 π fm t ) (3-1)
• Carrier signal: vc = Vc sin (2 π fc t ) (3-2)
• Amplitude modulated signal: vAM = Vc + Vm sin (2 π fm t) (3-3)
The method to produce a usable AM signal is simply to combine the information signal (fm) and the carrier signal (fc) through a non-linear device, such as diodes, transistors or even electronic valves, which is called the mixer or sometimes known as modulator. A mixer normally has two inputs for modulating and carrier signals. In Figure 3.1 shown below, the mixer multiplies the two signals together and the output comprises a mixture of frequencies as follows:
• Components of original modulating and carrier signals, fm & fc.
• Components of the sum and difference of original signals and harmonics of original signals; (fc+fm) & (fc-fm), (2fc+fm) & (2fc-fm), (2fc+fm) & (2fc-fm) ……. etc.
• Harmonics of original signals, 2fm, 3fm, …., 2fc, 3fc …… etc.


In practical situation, a filter will be used to get rid of all the frequencies except (fc – fm), fc and (fc + fm), which we call AM Double-Sideband-Full-Carrier (DSBFC) waveform.

Figure 3.2a shows a very simple Amplitude Modulator constructed by a JFET. It operates with gate bias by R4 to the first non-linear region as shown in Figure 3.2c. The level of the modulating and carrier signals should be well adjusted to be operated within the non-linear region through out their maximum voltage swing to guarantee proper generation of undistorted AM wave. A parallel LC tank circuit is connected to it’s drain and tuned at resonant frequency of the carrier (fc) to remove unwanted frequency components except the carrier and the two side-bands. Figure 3.2b shows the input-modulating signal, the carrier signal and output AM waveform (DSBFC).
































3.2.1 AM ANALYSIS
With reference to the amplitude of an AM wave in equation 3-3. The instantaneous voltage, ec of the resulting amplitude modulated wave can be express by:
ec = vAM sin (2 π fc t ) = {Vc + Vm sin (2 π fm t)} sin (2 π fc t )}
= Vc {1+ m sin (2 π fm t)} sin (2 π fc t ) (3-4)
Where m = Vm/Vc is defined as the modulation index (3-5)

Equation (3-4) may be expanded by the relations of trigonometry of, sin x sin y = {cos(x -y) – cos(x+y)} / 2, to give
ec = Vc sin (2 π fc t ) + {mVc/2} [cos{2 π( fc - fm )t}
- cos{2 π( fc + fm )t}] (3-6)

Here, we see that the amplitude term has been replaced with the combination of the original amplitude plus the information signal. The amount of modulation depends on the amplitude of the information signal. This is usually expressed as a ratio of the maximum information signal to the amplitude of the carrier. We define the modulation index or depth of modulation m = Vm/Vc.

From the above equation (3-6) of the amplitude modulated signal contains three major components. The first is the original un-modulated carrier; the other two components produced are the two sidebands. The frequency of the lower side frequency is fc - fm, and the upper side frequency fc + fm. At this stage, the very important conclusion can be made is that the bandwidth required for successful transmission of an amplitude modulated signal is twice the frequency of the modulating signal.

BWAM = ( fc + fm ) - ( fc - fm ) = 2 fm (3-7)

The appearance of the amplitude-modulated wave is shown in Figure 3.3a for a 1KHz-modulating signal with the modulation index = 100%, i.e. amplitudes of the carrier and information are equal, modulating a 10KHz carrier signal. The related frequency spectrum may be represented in Figure 3.3b




From Figure 3.3b, which shows the frequency spectrum consisting of three discrete frequencies, they are the center frequency, i.e. the carrier, has highest amplitude, and the other two side frequencies disposed symmetrical about it with equal amplitude, 1/2 amplitude of carrier but never be allowed to exceed half the amplitude of the carrier, i.e. modulation index (m) must be <=1. Otherwise, distorted modulated wave will be resulted. In actual situation such as the AM broadcasting service, modulating signal is rarely a pure sinusoid (tone), but rather it is a complex mixture of pure sinusoids, such as voice or music, which superimposed with a series of sine waves simultaneously, the spectrum becomes filled with a wide variety of notes and their harmonics. Hence, the bandwidth required is twice the highest modulating frequency. A typical signal might look like as shown in Figure 3.4a and Figure 3.4b


For illustration purpose, the carrier is at 65 KHz and the information signal used to modulate has its own spectrum ranging from about 1KHz to 11 KHz. The spectrum has the usual appearance: a strong carrier signal is at the middle, and the two symmetrical sidebands. The bandwidth: BWAM = 2 fm, where fm is now the maximum modulating frequency of 11KHz.

3.2.2 POWER DISTRIBUTION IN AM WAVE
From equation (3-6), the total power in the modulated wave will be

Pt = Pc + PLSB + PUSB ........................... (3-8)
Pt = (Vc2 / 2R) + (VLSB2 / 2R) + (VUSB2 / 2R) .............................. (3-9)

Where: Pt: Total power of the AM wave Vc: Amplitude of carrier VLSB: Amplitude of LSB VUSB: Amplitude of USB R: The resistance (e.g. Antenna resistance)

Since VLSB = VUSB = mVC/2
PLSB = PUSB = (mVc/2)2 / 2R = m2Vc2 / 8R ..................... (3-10)

Substituting equation (3-10) into equation (3-8), we have, Pt = (Vc2/2R) + (m2Vc2/8R) + (m2Vc2/8R) = Pc + m2Pc/4 + m2Pc/4 = (1 + m2/2) Pc (3-11)

Equation (3-11) determines the total power of the AM wave, which is dependent on the value of the modulation index (m). It also anticipates that the maximum power in the AM wave is Pt = 1.5Pc when m=1. This is very important indication that the maximum power of the relevant RF amplifier must be capable of handling without introducing distortion. Also, we can also see that the carrier signal is very constantly and steadily dissipating power through out the transmission process, even there is no input signal. Hence, one of the simplest transmission method, double-side-band-suppressed-carrier (DSBSC) was devised to save power, which will be briefly described in next section.

3.2.3 DOUBLE-SIDEBAND-SUPPRESSED-CARRIER (DSBSC)
Equation 3-7 showed that when a carrier is amplitude modulated by a pure sine tone, the resulting AM wave consists of the original carrier signal and two frequencies. Experiments proved that the carrier of standard DSBFC (Double-sideband-full-carrier) AM conveys no information and all the information can be conveyed by use one or two of it’s sidebands only. The resulting signals will require less transmitted power or will occupy less bandwidth. There are many varieties of sideband transmission methods, they are: • Single-Sideband-No-Carrier (SSB) • Single-Sideband-Suppressed-Carrier (SSBSC) • Twin-Sideband-Suppressed Carrier, or Independent Sideband (ISB) • Double-Sideband-Suppressed-Carrier (DSBSC) DSBSC is the most simplest sideband transmission method that will be implemented in this project; brief analysis of this modulation method will be treated in some detail in this chapter. Recall the equations 3-10 and 3-11; carrier power stays constant in the entire modulation process; i.e. 100% to 66.7% of power is wasted for the modulation index (m) from 0 to 1 respectively. It is obvious; that removing the carrier from the standard AM (DSBFC) wave can save at least 66.7% of total power. However, due to the lack of carrier frequency, recovery of the original modulating information is highly dependent on the accuracy of the regenerated carrier in the receiver. Hence, in this type of DSBSC transmission method, the carrier will not be totally removed but suppressed to a desired level. The suppressed carrier can then be used at the receiver as a reference signal or so called pilot frequency for the AGC and AFC circuitries to regenerate a phase coherent carrier, demodulation of the information can then be achieved.

Figure 3.5 shows a simple circuit of a balanced modulator, which generates both sidebands but eliminates the carrier. This circuit is configured as a conventional push-pull amplifier but wisely feeding the carrier signal in-phase to both gates of Q1 & Q2 JFETs, the carrier is ultimately cancelled by the output transformer T3. However, the effect of the modulating signal is different, the modulating voltage is applied to the gates of Q1 & Q2 JFETs in 180o out of phase. Because of the non-linearity characteristics and the push-pull arrangement of Q1 & Q2, the transformer T2 combines the modulating frequency and both sideband frequencies. However, the modulating frequency is so low that is highly attenuated at the output of T3 and finally, only both sideband frequencies are reflected at the output of T3.

Figure 3.6 shows how a DSBSC wave is demodulated to the original modulating signal by the reinsertion of carrier, that the carrier used in the demodulation process is recovered from the already inserted pilot tone in the transmitted signal. After reinsertion of the carrier, the DSBSC is transformed into a normal DSBFC wave and then followed by a normal envelope detector and LPF used in AM receiver to reproduce the original modulating signal. Figure 3.6 Demodulation of DSBSC by insertion of carrier

3.4 FREQUENCY MODULATION
Frequency modulation uses the information signal, Vm(t) to vary the carrier frequency within some small range about its original value. Here are the three signals in mathematical form: • Modulating signal: Vm(t) = Vm cos (m t + ψ) (3-12) • Carrier signal: Vc(t) = Vc sin ((c t +θ) (3-13) • Instantaneous frequency f of the frequency modulated wave: f = fc (1 + Kv Vm cosmt) (3-14) Where Kv is proportionality constant.

A very simple circuit is as shown in Figure 3.7, which consists of an LC oscillating tank circuit generating a sine wave with frequency, fc. The capacitance of the tank circuit is systematically made variable by the modulating signal, i.e. the amount of capacitance change is proportional to the amplitude of the modulating signal. Since the capacitance of the tank circuit has the direct effect on the oscillating frequency, ultimately, the output frequency of the tank circuit is modulated by the modulating signal.

In Figure 3.8 shown, it represents the way how the modulating signal Vm(t) modulates the carrier signal Vc(t). The output FM signal VFM(t) is made oscillating about the carrier center frequency (fc), the amount of frequency deviation is proportional to the instantaneous voltage of the modulating signal, where the proportionality is equal to the sensitivity of the voltage controlled oscillator. It also shows that the frequency variation of the FM signal is identical to the variation with time of the modulating signal and the amplitude of the frequency-modulated wave remains constant at all times, which form one of the major advantages of FM that amplitude variation wouldn’t degrade the quality of the received signal. Figure 3.8 FM representation 3.4.1 FM ANALYSIS Consider a voltage controlled oscillator (VCO) such as shown in Figure 3.7 with a free running frequency of fc, a modulating signal voltage Vm(t) which causes the VCO to vary about its carrier frequency (fc) between the extremes of (fc –δ) and (fc +δ), where δ is the maximum amount of frequency deviation equal to the voltage of the modulating signal times the sensitivity of the VCO, Kv in term of Hz/V. The concerned output frequency modulated signal can be expressed in the three basic equations 3-12, 3-13 and 3-14 mentioned in section 3.4. Full mathematical analysis of the frequency modulation requires complex mathematical derivations and will not be covered in this report. Details of mathematical analysis can be obtained from many textbooks. From equation 3-14, the instantaneous frequency of a FM wave is: f = fc (1 + Kv Vm cosmt) or f = fc (1 + δ cosmt) (3-15) Where δ is the maximum frequency deviation The instantaneous amplitude of the frequency-modulated signal will be given by a formula of the form: VFM(t) = Vc sin [F(c, m)] = Vc sin (3-16) Where [F(c, m)] is some undetermined function, after solving the  by using complex integration method with respect to time, gives the instantaneous value of the FM voltage as: VFM(t) = Vc sin {mt + (δ/fm) sin mt} (3-17) Obviously, the modulated wave contains a time varying frequency analogous to amplitude modulation; the modulating index for FM is defined as: Modulating Index (mf) = δ/fm (3-18) Where fm is the maximum modulating frequency used. Thus equation 3-17 can be rewritten as: VFM(t) = Vc sin {mt + mf sin mt} (3-17) 3.4.2 FM bandwidth and required spectra In FM, both the modulation index and the modulating frequency affect the bandwidth. As the information is made stronger, the bandwidth also grows. Hence, determining the bandwidth for FM is very complex and the theoretical bandwidth required in FM transmission is infinite. In equation 3-17, it shows that the bandwidth is very dependent on the value of the modulation index (mf). In practice, Carson’s Rule gives an indication to the bandwidth required by an FM signal. Carson’s Rule states that the bandwidth (BW) in Hz is twice the sum of the maximum carrier frequency deviation and the maximum modulating signal frequency, i.e. BW ≈ 2 (δ + fm) or BW ≈ 2 fm (mf + 1) (3-18) In this project, the receiver unit is using the conventional FM broadcasting standard that the audio signal to be transmitted ranges from 15 to 15,000 Hz. If the FM system used a maximum modulating index (mf) = 5, then by Carson’s Rule, the resulting bandwidth is 180 kHz.

The FCC assigns stations 0.2 MHz apart to prevent overlapping signals as shown in Figure 3.9. If channel separation between each FM stations is 200KHz, then the total number of FM stations fills up the FM band is: 108 - 88 / .2 = 100 stations. Figure 3.9 FM Radio Channel Bandwidth However, maximum modulating frequency is 53KHz for FM stereo broadcasting, containing the following three signals, which will be briefly discussed in next chapter. • L + R signal in the range of 15 to 15,000 Hz. • 19 KHz pilot frequency. • L-R signal centered on a 38 KHz sub-carrier (which is suppressed) that ranges from 23 to 53 kHz. Hence, according to Carson’s Rule, the modulation index (mf) must be about 1.0 to keep the signal within the 200KHz bandwidth.


CHAPTER 4: BASIS OF STEREO SIGNAL ENCODING

As most of the household Audio-Visual electronic products are in stereophonic, to convey these type of signal through the RF link, some encoding and decoding strategies are crucial to reproduce the stereo sound or music. This involves transmitting and receiving two related audio signals, the ‘left’ and ‘right’ channel signals required to produce a stereo image. In principle, three different approaches, that can be used to convey stereo information. I will briefly describe them in turn.

4.1 INDEPENDENT CHANNEL METHOD
We can use two complete transmitter/receiver systems to send stereo sound as shown in Figure 4.1. but this impractical approach would lead to many disadvantages and drawbacks. Such as: • Twice the standard FM broadcast bandwidth, i.e. 400KHz • Incompatible with monophonic FM systems. • Requires special receivers with two independent tuners. • Transmission delay on both TX/RX systems due to the path diversity will destruct the coherence of “left” and “right” channel signals.

4.2 DIRECT APPROACH TO FM STEREO
The feasible approach is to transmit the left and right signals in the same channel frequency. The direct and intuitive approach is by sending the left channel, and then translate the right channel signal above the normal audio spectrum within the available bandwidth of the channel. This method can be illustrated as shown in Figure 4.2.

However, such system is not compatible with the existing mono-aural receivers, because the monophonic equipment would detect only the left channel. Furthermore, the percentage of modulation for the left channel must be reduced to provide headroom for equal representation of the right channel signals. Otherwise, unbalanced or distorted outputs will be resulted or the reduction of modulation would seriously downgrade the quality of mono-aural reception.

4.3 SUM & DIFFERENCE SCHEME OF MULTIPLEXING
In practice, we may consider the issue of compatibility with the mono system when introducing the new design of stereo system. Method for conveying of the stereo signal through the original limited bandwidth is applicable. Hence, stereo signal encoding is more effective and sensible to use the frequency division multiplexing (FDM), by which the stereo signal information can be transmitted in a specific frequency slot within the available bandwidth. Nowadays, the standard analogue FM stereo system, which has been adopted by commercial broadcasting stations and consumer RF stereo products, is illustrated in Figure 4.3.


The system uses sum & difference method and frequency division multiplexing (FDM) technique to combine the two left (L) and right (R) channel signals. The signals are first passed through low-pass filters, which only allow through frequencies up to 15 kHz with very sharp roll-off characteristic. The L and R signals are then added to produce a sum signal and subtracted one from the other to produce a difference signal. The sum is essentially a monophonic signal, which is what we called the downward compatibility and partly used for the proper reception by mono FM receivers. The difference signal is used to DSBSC modulate a 38 kHz sub-carrier.

The DSBSC output is added to the sum signal and the 19KHz pilot tone, the combination is then sent to the transmitter's FM modulator. A monophonic receiver can now ignore the stereo information simply by using a filter after its FM demodulator to block everything above 15 kHz. A stereo receiver has to have an additional circuit after the FM demodulator, which can detect and demodulate the DSBSC wave such as shown in Figure 3.6. Once it has done this, it has recovered the difference information and can recreate the left and right signals by adding/subtracting between the sum and difference signal, the process can be simply illustrated by the following mathematical equations: L = (Sum + Diff) / 2 = {(L + R) + (L – R)} / 2 (4-1) R = (Sum - Diff) / 2 = {(L + R) - (L – R)} / 2 (4-2) In section 3.2.3, we can see the advantages of DSBSC modulation and anticipates why this modulation method be used during this stereo encoding process. If the 38KHz carrier component were not suppressed, which is not only waste power i.e. contains twice the power of the two sidebands, but does not contain any information as well. If it remains in the modulating signal, a continuous but useless modulation of the main carrier would occur. This would largely reduce the modulation percentage of the main L+R modulation and downgrade of reception on monophonic receivers. Hence, by suppressing the 38KHz sub-carrier, a great saving in modulation percentage and power. Whereas, due to the absence of the carrier, demodulating a DSBSC signal can be difficult whose frequency & phase we need to perform demodulation. In the stereo system, this problem can be alleviated by including a 19 kHz pilot tone. This comes from a 19 kHz master oscillator at the transmitter to which the 38 kHz sub-carrier oscillator is phase locked. The 19 kHz pilot falls in a spectral region above the mono sum signal and below the DSBSC difference signal information. (The DSBSC signal extends ±15 kHz around 38 kHz since the input modulating signals are band limited to 15 kHz.). If a 19 kHz pilot tone is detected, stereo transmission can be recognized and used to control the frequency and phase of a 38 kHz oscillator in the receiver's stereo decoder. This can then demodulate the difference information and combine it with the mono (sum) signal to recover the stereo sound.


CHAPTER 5: BASIS OF PLL & FREQUENCY SYNTHESIS

5.1 BASIS OF PHASE LOCKED LOOP
Phase-Locked-Loop (PLL) is a well-established and very widely used technique in modern circuit design. Basically, a phase-locked-loop is a closed-loop controlled system, which generates a very stable output frequency by continuously monitoring and maintaining a constant phase relationship between a reference signal and an oscillator output signal.

Figure 5.1 shows a basic building block of a PLL. A phase frequency detector compares the phase of the voltage-controlled-oscillator (VCO) output frequency, fosc, with the phase of a reference signal, fref. A phase detector output pulse is generated in proportion to the phase difference. This pulse is then passing through a low-pass-filter and generating a DC voltage, which in turn is used to control the VCO. The output of the VCO, fosc, is fed back to the phase detector input for comparison, which in turn controls the VCO to minimize the phase difference. Finally, in the equilibrium and locked state, both the frequency and phase of reference and VCO are the same, i.e. fosc = fref and θosc = θref.

5.1.1 BRIEF ANALYSIS OF A PLL
A simple analysis of a PLL can be performed by using the linear model of negative feedback control system is as shown in Figure 5.2.

The parameters in Figure 5.2 are defined as follows: Kp: gain of the phase detector (V/rad) Kf: transfer function of loop filter (V/V) Vc: VCO control voltage Kv(s): gain of VCO (Hz/V) S: Laplace transform variable



The behavior of a PLL can be treated as similar as a feedback voltage amplifier in Figure 5.3.

Hence, by applying Mason’s rule, the closed-loop transfer function for a single- loop control system can be reduced to the following simple formula: Closed-loop transfer function = Output / Input = (Path gain from input to output) / (1 – loop gain) (5-1) Using a Laplace transform, the closed-loop transfer function of a PLL can be expressed as: θosc(s) /θref(s) = {Kp x Kf(s) x Kv(s)} / {1 + Kp x Kf(s) x Kv(s)} (5-2) The VCO transform gain, Kv is a function of time integral of frequency, the gain can be expressed as Kv(s) = Kv/s (5-3) From equation (5-1) and equation (5-2) θosc(s) /θref(s) = {Kp x Kf(s) x Kv} / {s + Kp x Kf(s) x Kv} (5-4) Equation (5-4) is the general transfer function for a PLL.

5.2 SYNTHESIZER ARCHITECTURE
Most RF transceivers design use frequency synthesizers to generate the highly accurate frequencies used for the transmitter carrier and receiver local oscillator. All synthesized RF equipment use a phase-locked-loop (PLL) circuit that described in section 5.1, to control the operating frequency. One of the most common frequency synthesizer architectures in RF transceiver applications is the indirect PLL synthesizer, it’s architecture is depicted in figure 5.4, can be used to generate a number of closely spaced RF output frequencies.

A frequency synthesizer consists of two oscillators (a reference oscillator and a VCO), a phase detector, loop filter, and a frequency -divider. The VCO is tuned by the phase-locked-loop to be a multiple of the reference frequency. Consider the operation of the PLL frequency synthesizer in Figure 5.4. A crystal oscillator is generally employed for the reference due to its low phase noise as well as its high accuracy, which insures good frequency matching between the transmitter and the receiver. When the phase-locked-loop is in equilibrium, the negative feedback loop forces the phase and frequency of the output to match that of the reference (operation principle refers to section 5.1). However, the output frequency is divided by N before comparison with the reference, its synthesized output frequency in lock state, fosc, is given by the relationship as follows: fosc = N x fref (5-5) Where N is the division ratio of the frequency divider. The frequency divider in the feedback loop is usually implemented as a programmable divider, achieving the output frequency to be in increments of the reference frequency. For example, suppose in a FM transmitter is operating from 88MHz to 108Mhz and the channels are to be spaced at 100KHz intervals. The reference frequency is again set to 100KHz in order to match with the channel spacing. With a programmable divider capable of dividing by 880 to 1080, output frequencies in the range of 88MHz to 108MHz can be synthesized with the selection of the appropriate divider ratio (88MHz = 880 x 100KHz, 88.1MHz = 881 x 100KHz, …………., 108MHz = 1080 x 100KHz).

5.2.1 BRIEF ANALYSIS OF A FREQUENCY SYNTHESIZER
A simple analysis of a frequency synthesizer can be performed using the linear model of negative feedback control system as shown in Figure 5.5.

The parameters in Figure 5.4 are defined as follows: Kp: gain of the phase detector (V/rad) Kf: transfer function of loop filter (V/V) Vc: VCO control voltage Kv(s): gain of VCO (Hz/V) S: Laplace transform variable N: division ratio Since the basic architecture of a frequency synthesizer is the same as a PLL described in section 5.1, except that a frequency divider be inserted in the feedback loop. Hence, the closed-loop transfer function of a PLL in equation 5-4 can be considered and the transfer function of a frequency synthesizer can be expressed by the following equation: θosc(s) /θref(s) = {Kp x Kf(s) x Kv} / {s + Kp x Kf(s) x Kv / N} (5-6) Which form the general equation of both PLL and frequency synthesizer’s transfer function. If the parameter N in equation 5-6 is set to 1, it becomes equation 5-4. The loop filter in a phase-locked-loop is used to compensate the negative feedback loop used in the system. It is also used to set the appropriate unity gain frequency and guarantee stability when the other parts of the system have been specified (phase detector gain, VCO gain, divider ratio, etc.).

The design of the loop filter is similar to the design of compensating networks in operational amplifier applications. One of the loop filter designing criteria is making low-frequency open loop gain in the system as high as possible, this helps the loop to reject noise and phase noise added within the loop. A typical passive loop filter is shown in figure 5.6.




CHAPTER 6: SYSTEM ARCHITECTURE DESIGN OVERVIEW


After the brief descriptions and understandings of fundamental concepts of how information signals are processed, transmitted and received, this chapter’s main intention is to discuss the concepts, methodology and the way to integrate the RF audio link.

The RF Stereo Audio Link contains three core functional modules, which contribute to the functionality of the system as a whole, they are: RF Remote Control TX Unit, RF Audio Link Base Unit and Receiver Unit. The Control Link and the Audio Link are operating in two distinct frequency bands, which can eliminate the possible interference between them. The following diagram, Figure 6.1 depicts the system setup configuration of this project.

The Control Link is operated in the AM 433.92MHz European Industrial-Scientific-Medical (ISM) frequency band. The application used in the RF wireless control is subject to the specific country regulations and restrictions. In Europe, this would be covered in European Telecommunications Standards Institute (ETSI) regulations, where wireless control systems are assigned to the 433.05MHz to 434.79MHz frequency range by EN300-200-1.

The Audio Link is operated in the FM broadcasting band 88MHz-108MHz, which is covered under Part-15 Sec. 15.239 of the FCC's rules for unlicensed operation on the extremely low powered transmitter. The reasons for the selection of the FM lie in their efficient use of transmitted power, low power modulation, and excellent noise immunity, by which the available 200KHz bandwidth achieves the high quality transmission of music and audio programme. For a full discussion of the signal processing issues and an analysis of the suitability of FM radio signals for use in RF transmission, see Gary M. Miller’s [1996] Modern Electronic Communication and William Schweber [1999] Electronic Communication Systems.

The Control Link comprises the Hand-held Remote Control TX Unit and the Remote Control RX Unit, which carries the control information that represents the source input button pressed by the remote user. Upon receipt of this control signals by the RF Control RX Unit inside the RF Audio Link Base Unit, the control information will then be demodulated and ultimately decoded to drive the source selection logic. The intended source signal will then be routed to the Stereo Signal Encoder, followed by frequency modulation process and ultimately radiated out through RF transmitter.

The Audio Link’s transmitter comprises the Stereo Signal Encoder (MPX), an 88MHz to 108MHz VHF Synthesizer and a RF amplifier interconnected as shown in Figure 6.2. The main function of the MPX is for signal level conditioning, pre-emphasis, band-pass filtering and encoding for stereo FM transmission, which is a standard method used in nowadays FM stereo broadcast transmission. The VHF Synthesizer uses phase-lock-loop method to generate a stable carrier frequency with 100KHz frequency step starting from 88MHz to 108MHz, i.e. ~200 channels can be synthesized. The process of frequency modulation is also achieved by adding the MPX signal after the loop filter of the VHF Synthesizer, which acts to directly modulate the VCO frequency. The modulated carrier will further be amplified by an RF amplifier and radiated through the antenna.

Lastly, the Receiver Unit is to detect and convert the information carried by electromagnetic waves through the audio link to audio and then conveyed to the ear by a speaker. It is actually a common FM stereo receiver, which may be in any form, e.g. hand-held, portable, headphone or even desktop model that exactly matches the user’s own preference. The high availability and low cost of these FM receivers provides benefits and features of compatibility and portability of this RF Stereo Audio Link system. After devising and finalizing the whole system configuration and design strategies for this RF stereo Audio Link, the next and crucial stage is to design circuits and integrate them into a feasible and workable entity. Furthermore, the development of new circuits and applications for this project is an interesting and often a challenging task. Since the work presented here is specifically focused on designing a feasible, reliable and high fidelity RF Audio Link, which is mostly based on some already established theories and newly devised implementation methods. The following chapters present detailed circuit designs, modifications, specifications and discussion of each components and modules.


CHAPTER 7 : CONTROL LINK DESIGN & IMPLEMENTATION

The previous sections in Chapter 6 briefed about system level architecture and how the system is implementing its remote control features, audio and RF signal processing. This chapter details the Control Link’s transmit and receive signal chain.

7.1 HAND-HELD REMOTE CONTROL TX UNIT
Figure 7.1 is the block diagram of the Hand-held Remote Control TX Unit, which is made up by four interconnected parts; they are transmitter module and antenna, control information encoder, push buttons and de-bouncing circuit, and finally the power supply. All components are housed in a 12cm x 6cm x 2.5cm plastic box and is shown in Photo 7.1.









































7.1.1 433.92MHz AM Transmitter
The RF transmitter section employs the Jablotron TX-3 433.92MHz AM Hybrid Transmitter Module, which provides a complete RF transmitter to transmit data up to 3.5KBPS. Figure 7.2 illustrates the circuit diagram of TX-3, the input-modulating signal and output AM waveform. This simple amplitude modulator is constructed by a NPN transistor and operates with base biased by R3 to the first non-linear region and the reasons are as described in section 3.2.

The base-band control data and the 433.92MHz carrier are being summed at the base of Q1. A parallel LC tank circuit is connected to its collector, and tuned at resonant frequency of the carrier to remove unwanted frequency components except the carrier and the two sidebands. The antenna is a loop antenna that has already been etched a closed-loop of copper tracks on a PCB of TX-3.

7.1.2 CONTROL INFORMATION ENCODER
There is little that can be done to the internal circuit of TX-3 besides building a control circuit that can send a serial control bit stream to the data input. One way to accomplish the control function is implemented by microprocessor at the control transmitter and receiver that are programmed with bit synchronization, error detection, correction and data recovery algorithms. However, the control function within my project is just for selecting input sources. Hence a much simpler way is to use an encoder IC at the transmitter and a decoder IC at the receiver. These ICs automatically generate and decode serial codes under their built-in data transmission/reception algorithms for filtering out or ignoring those unwanted signals to prevent false data from being received.

The complete circuit diagram is shown in Figure 7.3. Holtek’s HT12E is used as the control signal encoder, which provides 8 bits address inputs with 4 data inputs, i.e. 256 possible addresses that can be programmed. This IC requires only a few external components to put it into operation, which not only simplifies the design and reduces the cost as well. In this design, all address bits are connected to ground to further simplify the design and reduce components. 4 push buttons are installed between the 4 data-input pins and ground, which represents the 4 different source inputs in the Remote Control RX Unit. Once any of the push-buttons are being pressed, Transmit-Enable pin (TE) will be tied low and 4 groups of pulses containing address and data information is generated serially from data out pin 17 and the transmission of the four groups will continue to repeat the same address and data information as long as the push-button is pressed. The serially data will directly be routed to the TX-3 transmitter module and amplitude modulates the 433.92MHz carrier. Finally, the information will be radiated through the antenna.

The rate at which these address/data pulses are generated is controlled by an on-chip oscillator. The frequency of the oscillator is controlled by the 750K resistor R1 connected on pins 15 & 16 to set the internal clock about 4.2KHz and the transmit bit rate is 1.4Kbits/Sec. It takes about 68ms to send four groups of encoded pulses, i.e. the transmission rate is about 58 Ctl-words/sec. This low data rate is more than enough for this type of control purpose. The one and zero representation and the Address/Data bit waveform for the HT12E are shown as below Figure 7.4a & 7.4b. Figure 7.4a “1” & “0” Representation Figure 7.4b Address/Data Group









7.1.3 DEBOUNCING MECHANISM

No need to say, mechanical switches used in this remote control unit will have contact bounce, which may gives a series of erroneous pulse trigger signals on the data and transmit enable (TE) input such as shown in Figure 7.5. Unexpected input selection will be resulted.

Simple single-shot debouncing circuit is designed to eliminate this type of contact bounce. D5 to D8 and R2 configured as a simple Diode AND gate, which provides trigger input to the single-shot generator formed by R13 & C5 whenever any one of the buttons SWA to SWD is being pressed. Empirical results show that the average duration for pressing a remote control button is around 1 second; hence the time constant () set by the values of R3 & C5 should be less than 1 second in order to neglect the erroneous pulses before releasing the button; hence 700ms is chosen (where R3 = 15K & C5 = 47uf).

Figure 7.6 shows the resulting circuit waveforms and the voltage at the R3 – C5 junction, which goes from low to high. This ends the circuit’s unstable state after a time interval “T” where “T” is some function of the time based in R3 and C5. Generally, T = 0.7RC. During the active state of transmit enable (TE), about 28 groups of address/data information will be sent; it is far enough for the receiver to validate the address/data.


7.1.4 POWER SUPPLY
To increase the portability and decrease the standby power consumption of this unit, it is powered by a common 9V battery and controlled by a micro-switch respectively. As this unit is operated in UHF, RFI between components and modules must be well considered. In order to eliminate the interference and false operations, power supply lines between RF modules and control modules are isolated by a RFI filter, which is configured by C1, C2, L1 C3 & C4 to form a C-L-C (π) filter.

7.2 REMOTE CONTROL RX UNIT
Figure 7.7 is the block diagram of the Remote Control RX Unit, which is made up by four interconnected parts; they are Receiver module and antenna, control information decoder, input selection logic, and finally the power supply. The components layout is shown in Photo 7.3.














7.2.1 433.92MHz AM RECEIVER
The RF receiver section of the Remote Control RX Unit employs the Jablotron Hybrid AM Receiver Module RX-3, which operates at the carrier frequency of 433.92MHz and provides a complete RF receiver. It can be used to receive data up to 4KBPS. Basically, it is configured in a standard AM super-heterodyne receiver and its circuit diagram is shown in Figure 7.8. There is also little that can be done to the internal circuit besides building a decoder circuit that can decode the serial control bit stream to control the associated input selection logic.

During the construction and experimental process of this UHF receiver, some constraints and obstacles that were found to be very influencing to the stability and operation range of this receiver unit. When building this circuit a PCB can be used as long as lead lengths are not excessive. Since this is an RF circuit, layout of the design is critical and coupling of input to output leads should be avoided. It is found that putting the receiver module too close from the rest of the electronics and insufficient ground plane will lead to a great deal of noise, interference and spurious spikes at receiver output and will totally destroy the performance. Well-regulated power supply, decoupling and filtering are also very important to the operation range.

Finally, the receiver module is isolated from the decoder and selection logic PCB and fabricated with a large ground plane to eliminate interference. External regulated power supply is routed to the receiver module with high frequency type of decoupling capacitors directly connected between module’s power leads and ground plane. Under this special construction arrangement, noise, interference and operation range are improved significantly. The placement and interconnection of receiver module and the decoder & selection logic is shown as Photo 7.4a & 7.4b.



















7.2.2 CONTROL INFORMATION DECODER
The completed circuit diagram for Remote Control RX Unit is shown in Figure 7.9. Holtek’s HT12D is used as the control signal decoder, which provides 8 bits address with 4 latched data outputs, i.e. 256 possible addresses that can be programmed. In this design, all address bits are connected to ground to match with the HT12E’s setting. The frequency of the on-chip oscillator is controlled by the 33K resistor R5 connected on pins 15 & 16 to set the internal clock about 210KHz, which is about 50*fosc(HT12E) = 50 x 4.2KHz and suggested by manufacturer’s operation manual for proper decoding.

HT12D does not provide any error correction. It just accepts valid data and rejects corrupted data. This means that a control signal that is corrupted at any means will not be output and must be resent again. The algorithm that HT12D used to validate the information that being received is to check the received data group for three consecutive times before activating the output pins. This is usually not a problem since a large number of repetitive data groups will be sent before the button is released (refer to section 7.1.3). HT12D provides 4 latched type data output pins whose data remains unchanged until new & valid data are received. The 4 parallel data will then be routed to the Input Selection Logic configured by 74HC04 and transistors to drive the associated relay and achieves the input source selection.

7.2.3 INPUT SELECTION LOGIC
The data output pins of HT12D is active low, in-order to enhance the current sinking capability to drive the associated relay, the construction of the Input Selection Logic is very straight forward by connecting the HT12D data output pins to a 74HC04 inverter and then drive the corresponding reed-relays by four NPN transistors. Four LED indicators are also connected for the indication of the selected input channel.






















7.2.4 POWER SUPPLY
As I mentioned in section 7.2.1, purity of power supply is very crucial to the success of the RF and control performance. In order to eliminate the interference and false operations, the Remote Control RX Unit’s power input is coming from a +12V DC regulator and then followed by a on-board UA7805 +5V regulator to further improve the ripple rejection. Supply lines between RF modules and control modules are isolated by a RFI filter, which is configured in C-L-C (π) filter. This design confirms the amazing effect to the operation range of the receiver.


CHAPTER 8: AUDIO LINK DESIGN & IMPLEMENTATION

This chapter details the transmit signal chain of the audio link. It includes detailed block diagrams and descriptions from input source signal to the antenna.

Figure 8.1 shows the system level functionality as a signal to be transmitted propagates from the stereo audio source (Left / Right Channel) through the Stereo Signal Encoder. The raw audio signals will undergo the process of level conditioning and 15KHz low-pass filtering, and multiplexed as a composite spectrum containing the left & right channel information. Following the stereo encoding stage, the transmit signal is sent to the VHF synthesizer board and frequency modulates the VHF carrier. The frequency-modulated signal is then sent to the RF amplifier stage and finally transmitted through the antenna as electromagnetic wave.

Photo 8.1 shows the transmitter’s internal layout. Pilot & Subcarrier Generator Remote Control Receiver Control Link Antenna VCO & RF Amplifier Remote CTL Decoder and Control Logic Audio Link Antenna Audio Input Jacks Reference Frequency, Frequency Divider & Loop Filter Stereo Signal Encoder.

8.1 PILOT & SUBCARRIER GENERATOR
An accurate and stable 19KHz Pilot and 38KHz Sub-carrier is very crucial in stereo signal encoding and decoding process. The L-R signal is mixed in a balanced modulator with a 38KHz sub-carrier to produce an amplitude modulated double sideband suppressed carrier signal, in which contains the stereo signal information. 19KHz pilot will be injected in the stereo composite signal and is used for the validation and detection of the stereo transmission in a stereo FM receiver, which also used for the recovery of 38KHz sub-carrier to accurately extract the L-R signal during decoding of the stereo signals. Normally, the 38KHz signal is derived from the same source as the 19KHz pilot tone in-order to maintain their phase coherency. Two typical methods can be used to generate a stable pilot and sub-carrier, they are:

Direct from crystal oscillator Method (Figure 8.2): Low tolerance 38KHz or any multiple of 38KHz crystal oscillator followed by a series of divider to generate 38KHz and 19KHz signal is the most simple and straightforward implementation method. However, these types of crystal are very rare in the market and must be tailor made from the manufacturer.

Phase Locked Loop Method: The most common use of PLL is in frequency synthesizers as shown in the Figure 8.3, any frequency (fo) can be synthesized from a single stable reference frequency (fr). This simply requires the use of a proper ratio divider (N) in the feedback loop of a PLL, i.e. fo = N x fr. This combines the ease of tuning and wide deviation of a VCO with the frequency stability of a crystal oscillator. However, complicated design, analysis procedures and consideration in the trade-offs between step resolution, lockup time, capture and lock range, sideband noise and the stability of the VCO are involved.

Figure 8.4 the building blocks of the Pilot & Sub-carrier Generator using Phase Locked Loop Method. Figure 8.4 Pilot & Sub-carrier Generator In the block diagram, there are four main parts, which make up the Pilot & Sub-carrier Generator by using a typical PLL configuration. These are a 4.096 MHz crystal to provide 100Hz reference frequency, 38KHz PLL and VCO, divide-by-two counter to provide 19KHz pilot frequency and the 19KHz & 38KHz resonant filters to remove harmonics and produce pure sine waves.

74HCT4046 acts as both the Phase Comparator and Voltage Control Oscillator, the Loop Filter as shown in Figure 8.5 is a simple passive second-order low-pass-filter for it’s simplicity, ideal behaviors for fast loop, stable and phase characteristic. The optimum design for the loop filter and the VCO is designed by using the freeware – 74HCT4046 PLL design program from Philips.

The following Table 8.1 shows the calculated results and characteristics of the VCO and loop filter using 74HCT4046 PLL design software.














The low tolerance 4.096MHz crystal oscillator is selected for ease of division by using a simple 74HC4060 ripple counter; 1KHz signal is generated after divided by 12-stages binary counter and followed by 74HCT390 decimal counter to provide a 100Hz reference clock as shown in Figure 8.7.

The particular connection for the 74HCT390 is configured as divide-by-5 then followed by divide-by-2 (Bi-quinary Count Sequence) to provide a 50% duty cycle square wave, and not vice versa (BCD count sequence). The output waveforms for both connections are illustrated as following Figure 8.8. If the input signals to the phase comparator have a duty cycle, which is far removed from 50% like BCD count sequence, then the response will be asymmetrical, resulting in a reduction in the operating range and impaired performance in a phase-lock application.

The 74HCT4046’s internal phase comparator compares the phase of the 100Hz reference signal with the phase of VCO output signal (refer to Figure 8.6) after dividing by a divided-by-380, which is configured by U4A, U4B, U4C, U4D and U5 as shown in Figure 8.9.

Then converts any difference between the phases into an error voltage signal. This signal is then applied to the loop filter, and the filtered output signal is used to modulate the 74HCT4046’s internal VCO. When the PLL is in "lock", the output of the VCO is phase-locked to the input signal and changes in input signal phase will be developed at the VCO output. The 38KHz will be further divided-by-2 by a 74HCT74 D-type flip-flop; a phase coherent 19KHz signal is then generated.

Both 50% duty-cycle square waves will pass through their own parallel resonant LC filter as shown in Figure 8.10b, which is equivalent to the standard RLC parallel filter circuit (Figure 8.10a), except that the equivalent parallel resistor is equal to the resistors Rs and the input impedance Rin of the following stage in parallel. Since the input impedance of the following stage is very high, ultimately, the equivalent parallel resistor of the tune circuit is approximately equal to Rs.

The pilot and subcarrier waveforms before and after the RLC filter and their phase relationships are shown in Figure 8.10c. Bw = fo/2Q = 1/4RC (Q = 2foRC) Bw = 1/4CRs//Rin (Q = 2foCRs//Rin) Figure 8.10a Standard RLC filter.

The filters are designed and tuned in the center frequencies of 38KHz and 19KHz respectively to filter the high frequency harmonics. The resonant frequency is given by fo = 1/2(LC)-1/2 and the resultant bandwidth (Bw) of the filter is equal to fo/2Q = 1/4RC where Q = 2foRC.



Table 8.2 tabulates the calculation of the components used in the 19KHz & 38KHz bandpass filter and the circuit arrangement for generating these pure sine waves is shown in Figure 8.11. Table 8.2 fo = 1/2(LC)-1/2 19KHz band-pass filter 38KHz band-pass filter Selected (L) 47mH 10mH Calculated (C) 1492PF 1754PF Practical (C) 1490PF 1750PF Selected (R) 91K 24K Q-Factor (Q) 16.2 10 Band-width (Bw) 1843Hz 5952Hz.

The 38KHz sub-carrier and the 19KHz pilot will then be pure sine waves and buffered by AD712. The output waveform and frequency of these pilot and sub-carrier are shown in Photo 8.2a and 8.2b.



















The complete circuit diagram and module of Pilot & Sub-carrier Generator are shown as following Figure 8.12 and Photo 8.3.


























8.2 STEREO SIGNAL ENCODER (MPX)

The function of this Stereo Signal Encoder (MPX) is accomplished by the building blocks shown in Figure 8.13. The Stereo Signal Encoder uses frequency division multiplexing technique to combine the Left/Right signals. The signals are first buffered and emphasized with a time constant (λ) equal to 75us, then passed through a 19KHz notch filter, which is implemented by a parallel LC resonant circuit to prevent the pilot frequency to beat with the audio signals. The L and R signals are then passed through a 15KHz LPF and added to produce a sum signal (L+R) and subtracted one from the other to produce a difference signal. The sum is essentially a monophonic signal, which is what we would send for playing through a single loudspeaker. The difference signal (L-R) is used to DSBSC (Double Side Band Suppressed Carrier) method to modulate a 38 kHz sub-carrier.

The DSBSC output is added to the sum (mono) signal and the combination is sent on the transmitter's FM modulator. A monophonic receiver can now ignore the stereo information simply by using a filter after its FM demodulator to block everything above 15 kHz. A stereo receiver has to have an additional circuit after the FM demodulator, which can detect and demodulate the DSBSC wave. Once it has done this it has recovered the difference information and can recreate the left and right signals by using an adder and subtractor circuit as Figure 8.14.

The generation of the Left/Right stereo signal is implemented mathematically as: L = { (L– R) + (L+ R) } / 2 (8-1) R = { (L+R) – (L–R) } / 2 (8-2) Whereas, demodulating a DSBSC signal can be difficult due to the absence of the carrier whose frequency & phase we need to perform demodulation. In the stereo system this problem is dealt with by including in the transmit signal a 19 kHz pilot tone, which is used for the validation of a stereo transmission and the recovery of the 38KHz sub-carrier for decoding stereo signals. The composite signal spectrum is illustrated Figure 8.15.

The Stereo Signal Encoder is one of the core parts of the RF link, which governs the functionality and the quality of the audio image reproduction. The following section gives a complete picture of the implementation method.

8.2.1 INPUT SIGNAL CONDITIONING
Figure 8.16 and Figure 8.17 show the actually circuit design of how the raw L/R channel signals are conditioned through their own respective signal chain. The signal first passes through an RLC filter that removes any stray RF from the audio inputs. The 10K potentiometers adjust the audio input level for each channel during normal operation. U1A and U2A buffer the input signals and provide constant impedance for the following pre-emphasis circuit. The 75K resistor in parallel with the 1nF capacitor provides pre-emphasis. U1B and U2B amplify the signals to provide enough drive for the following 15KHz low-pass circuit.

The amount of pre-emphasis is referred to the U.S FM broadcasting standard at 75us. A 75us pre-emphasis corresponds to a frequency response curve that is 3dB up at the frequency whose time constant RC is 75us and continues rising in the rate of 6dB/Oct. The frequency is given by f = 1/2RC and is therefore, 2120Hz. For the design of the pre-emphasis RC network, the chosen values for the parallel resistor and the capacitor are arbitrary, in which the commonly available values 75K ohm and 1nf are perfectly matching the standard.

Figure 8.18 shows the 75us pre-emphasis and de-emphasis curves. In Figure 8.17, the pre-emphasized signal undergoes a 19KHz notch filter to remove unwanted 19KHz frequency component. The 10mH and 7nF parallel resonant circuit with the resonant frequency 19KHz (fo =1/2(LC)1/2 = 1/2(10mH x 7nf)1/2 = 19KHz), is used to notch out any 19KHz signals that may come in from the audio input. This is necessary to eliminate beat frequencies that would be caused by combining 19khz with the pilot frequency. The filtered signal then continues going down the 8th order 15KHz Butterworth Low Pass Filter, Figure 8.17.

The purpose of this stage is to filter out the undesirable high-frequency components above 15KHz in the L/R channels. The low-pass filter is designed by using the Maxim MAX274, which is an 8th-order active filter. It is selected primarily because its independently cascadable 2nd-order sections provide maximum flexibility in the filter order, and it can implement several filter responses (Butterworth, Bessel, Chebyshev, and Elliptic). For reliable and high fidelity audio performance, minimum distortion to the waveform was a top priority. This means that good linear phase delay/response and a flat pass-band are vital. Using the DOS-based design program included with the MAX274, several classic implementations were examined and selected the 8th-order Butterworth filter based on the aforementioned criteria. The final design yielded a –3 dB pass-band of 15KHz and –48dB/Octave rate of attenuation. In selecting the low-pass filter, several options were considered, with the Maxim MAX274 active filter ultimately selected. The decision on the classic type of filter used relied on a comparison of the phase response and pass-band characteristics of the filter. Table 8.3 summarizes the different filter implementations achievable with the MAX274.

Based on the minimum wave distortion criterion discussed before, Chebyshev and Elliptic are ruled out because of the ripple pass-band. Butterworth was selected over Bessel for its flat pass band and steeper attenuation. Finally, EV-274 filter design program showed that an eighth-order Butterworth LPF provides the most optimum characteristic within the frequency range of interest.

The actual internal circuit and external components for each filter part of the MAX274 is determined by using EV-274 design program are shown in Figure 8.19a, b, c and d. The comparisons for the 8th order 15KHz LPF design among Butterworh, Bessel and Chebyshev by using MAX274 are shown Figure 8.20a, b and c respectively.

















8.2.2 Balanced Modulator and the Stereo Signal Matrix
The heart of the Stereo Signal Encoder comprises the Balanced Modulator and the Stereo Signal Matrix shown in Figure 8.21 and Figure 8.22 respectively.

The design is rather straightforward, U7B is a mixer circuit that sums Left, Right, the 19Khz tone, and the 38Khz modulated signal. The output of U7B is sent through an RF filter that keeps RF from nearby transmitters from getting back into the circuit. U5A is a differential circuit that takes Left and Right and produces the L-R difference signal. The L-R signal is fed into one of the two signal inputs, pin 4 of the LM1496N balanced modulator IC, U6. The 38Khz sine wave signal is fed into the other input pin 10 of U6. The amplitude modulated 38Khz signal shows up on pin 12 of U7. U7A is used as a signal buffer for the amplitude modulated 38Khz signal and the 10K pot is used to set the 38Khz mix level. The R61 20K pot on U6 is used to null out the 38Khz carrier on the balanced mixer. The 8V zener diode is used as a voltage regulator to turn the -12v supply into -8V for U6.

The following figures show the ideal output signal waveforms of the Stereo Signal Encoder (MPX), in which the 38KHz sub-carrier is totally suppressed and the 19KHz pilot is switched off for illustration purposes. In Figure 8.23a, two identical sinewaves are input to the Left and Right channel input, the output from MPX is exactly the sum of the Left and Right channel signal. However, 38KHz suppressed carrier amplitude modulation waveform with a flatline exists in between alternate halves of the envelope is shown in Figure 8.23b, when only LEFT channel signal presents. The flatter the line, the more precisely the LEFT-RIGHT signal will cancel the LEFT+RIGHT signal, i.e. the better the channel separation can be achieved. Hence, it is one of the methods in checking the channel separation performance of this audio link. No Right Signal Figure 8.23a MPX Output signal with two identical channel signals Figure 8.23b MPX Output signal with only right channel signal.

The completed module and components layout of the Stereo Signal Encoder (MPX) is shown in Photo 8.4 MPX O/P Filter L/CH 15KHz LPF Stereo Signal Matrix L/CH 19KHz Notch Filter I/P Amp & 75us network Balanced Modulator R/CH 15KHz LPF L-R Subtractor R/CH 19KHz Notch Filter.

8.3 VHF SYNTHESIZER
The VHF Synthesizer of this project is designed and constructed using a typical phase-locked loop, is depicted in following Figure 8.24.

Its purpose is to allow the phase of the synthesized output frequency to precisely track the phase of reference frequency (Fr) 12.5KHz. The phase detector compares the phase of the reference signal to the phase of a divided-down VCO output signal, and converts any difference between the phases into an error voltage signal. This signal is then applied to the loop filter, and the filtered output signal is used to control the VCO. When the PLL is in "lock", the output of the VCO is phase-locked to the reference frequency and changes in input signal phase will be developed at the VCO output. The output frequency (Fo) can be obtained by the following equation and is directly proportional to the value N of the divider. Fo = Fr x P x N (8-3) Where: Fr = 12.5KHz, P = 8 the Prescaler, and N is the value of programmable divider in this case. Fo = N x 100KHz (8-4) Hence, the frequency step size is 100KHz. The VHF Synthesizer not only generates a stable frequency but also very important and useful in frequency modulation, through which the combination of controllable modulation with a highly stable and adjustable carrier frequency can be achieved. Modulation can be applied to different points within the PLL for implementing frequency modulation. Two basic arrangement of a PLL frequency modulator can be implemented by adding a modulating signal either before at the VCO input or at the master oscillator. In this design, modulation is applied to the input of the VCO (refer to Figure 8.24) for its simplicity and higher sensitivity or frequency variation to the modulating signal. When modulation is applied to the input of the VCO, the modulation input from the Stereo Signal Encoder (MPX) is summed with the loop filter output and the composite signal is then used to the VCO. High-frequency content above the loop filter bandwidth of the modulating signal is developed at the VCO output. Low-frequency content that falls within the loop filter bandwidth is compensated by the PLL and is not present at the output of the VCO. This gives the PLL a high-pass characteristic with respect to the modulating signal, and modulation sensitivity is reduced at lower frequencies.

Figure 8.25 & Figure 8.26 are the completed schematic diagrams of the VHF Synthesizer in this project.


























8.3.1 Reference Frequency
The low tolerance 6.4MHz crystal oscillator is selected for ease of division by using a simple 74HC4060 ripple counter; 25KHz signal is generated after divided by 8-stages binary counter and followed by 74HCT390 decimal counter to provide a 12.5KHz reference clock as shown in Figure 8.27.

The 12.5KHz selected reference frequency has it’s purposes; first of all 100KHz step frequency can be easily obtained by the multiple of 8, which is implemented by two stages of 74ALS74 and one stage of 74HCT390 prescalers (Figure 8.25), i.e. 100KHz = 12.5KHz x8. Ease for the suppression of 12.5KHz ripple through the loop-filter is another reason. Figure 8.27 12.5KHz Reference Frequency

8.3.2 Programmable Counter
74HCT4059 programmable counter is selected as the core of the dividing counter in the feedback loop. The counter can be programmed by four sets of BCD dip-switches (Table 8.4), which is designed as the direct read-out configuration that the BCD reading shows the exact carrier frequency, such as: 0881 means 88.1MHz and 1073 means 107.3MHz, etc. Figure 8.28 divide-by-8 Prescaler and Programmable Divider.

However, this counter can only be operated in the maximum counting rate of 30MHz. Hence, a high speed pre-scaler must be introduced to step down the VHF to below 30MHz in order that the components inside the loop can be functioned normally within the synthesized frequency range. High speed D-type F-F 74ASL74 was chosen for this purpose since the maximum counting rate for this chip is 125MHz.



8.3.3 Voltage Controlled Oscillator
For Voltage Controlled Oscillator works in VHF region, Colpitts (split capacitor configuration) or Hartley oscillator (split inductor configuration) may be deployed.

Figure 8.29a & 8.29b show their configurations and their respective formulas for frequency of oscillation. fo= 1/2π{L(C1xC2)/(C1+C2)}-1/2 fo =1/2π{(L1 + L2)C}-1/2 .

The Colpitts oscillator will exhibit a smaller tuning range since the fixed feedback capacitors limit variable capacitance range; however, the Colpitts has good frequency stability with proper components. On the contrary, the Hartley will tune a very large range since all of the capacitance is variable. In the consideration of a wide-band frequency modulation of the composite stereo signal input with linear characteristic and a wide operating carrier frequency, this is achieved by using a lightly loaded high stability half frequency push-pull Hartley oscillator for generating the RF signal.

The output of the Hartley oscillator is configured as a frequency doubler using a push-push arrangement as shown in Figure 8.30a.

fo = 1/2π(LC)-1/2 where L = L1 +L2 + L3 + L4 fc = 1/2π{LcCc}-1/2

where fc = 2fo

Both transistors Q1 and Q2 are operated in Class-C push-pull configuration. Anti-phased signals with frequency fo feedback from indictors L2 and L3 will drive the base of Q1 & Q2, current pulses which are less than half cycle will then be added at the collectors of the transistors. After the pulsating current with fundamental frequency of 2fo passing through the LC tank circuit, a sine wave with 2fo will be generated.

The operation of this Hartley Frequency Doubler is illustrated as Figure 8.30b.

The capacitance C of the Hartley Frequency Doubler in Figure 8.30a is the main control mechanism for the oscillating frequency. Normally, it is made variable in accordance with the control voltage. The simplest and most practical way to accomplish the variable capacitance effect is to use varactor diode; the terminal capacitance of this device is made inversely proportional to the applied reverse biased voltage and its typical characteristic is shown as below Figure 8.31.

In practical application of varactor diodes in tune circuits, two different modes of connection are as shown in Figure 8.32a and 8.32b.

In Figure 8.32a, single varactor diode is connected in series with a capacitor C, which is used to block the DC from the control voltage. The choice of varactor in this circuit mainly depends on the tuning range. However, particular attention has to be made when the control voltage approaches 0V, as this may introduce non-linearity and poor Q-factor. Another modified approach, which is used in this project, uses two varactor diodes connected in back-to-back configuration as in Figure 8.32b. It provides a lower effective diode capacitance compared with Figure 8.32a. However, the major advantages it provides is to prevent RF rectification at control voltage approaching 0V, hence reduces the chance of producing inter-modulation products and simplifying the bias requirements. In Figure 8.30a, the Push-Pull Hartley oscillator’s tank circuit is comprised of one variable capacitor and an inductor, which can be comprised of combinations of some fixed, semi-variable capacitors, varactor diodes and some series inductors, and as a whole to determine the required frequency of oscillation. The following Table 8.5 tabulates the required components value of the Hartley oscillators.

 fosc = 1/2(LC)1/2 Minimum Oscillation Frequency (fmin) 43MHz Maximum Oscillation Frequency (fmax) 55MHz Impedance of the Inductor (Zo) at fc = (fmin x fmax)1/2 = 48.6MHz 50 Ohm Required Inductance L {L = Zo / (2 fc)} 164nH Minimum Capacitance Cmin {fmin = 1/2(LCmin)1/2} 85.6pf Maximum Capacitance Cmax {fmax = 1/2(LCmax)1/2} 52.3pf In Figure 8.34, the final circuit design of the VHF Synthesizer shows how two varactor diodes KV1310 are connected in parallel to obtain the required capacitance for achieving the tuning range of this VCO. The following Figure 8.33 is the tuning characteristic of KV1310.

8.3.4 Frequency Modulation within the synthesizer
Two different control signals are involved in controlling the VCO center frequency and to achieve the frequency modulation process; they are the control voltages from the loop filter and the MPX. The combined effect of modulating the VCO can be accomplished by two different methods as shown in Figure 8.35a and Figure 8.35b. (1) combining the control voltages at the junction of varactor diode pair and (2) control voltages from loop filter and MPX have their own varactor diode pair.

• Single varactor pair Method (Figure 8.35a) – This method is simple and stable for fixed frequency of operation as long as the control voltage is varying in a small amount. However, at the MPX signal chain, the DC blocking capacitor C1 and the resistor R1 to prevent further loading effects on the VCO tune circuit. The combined effect of C1 and R1 also adds the extra time-constant to the synthesizer’s loop-filter, this will increase the lock-up time of the synthesizer and eventually over correcting or unlocked condition will be resulted. Uneven modulation depth at any given control voltage is another problem for this configuration. When the control voltage from the loop-filter is low, large terminal capacitance change for small voltages from MPX, i.e. high frequency deviation level. Whereas, frequency deviation will drops dramatically if the loop-filter control voltage is high. The symptoms are illustrated in Figure 8.36.

• Dual varactor diode pair Method (Figure 8.35b) – The disadvantages mentioned above can be improved by adding an extra varactor diode pair. Control voltage from synthesizer’s loop-filter is applied to the varactor diode pair ZD3 and ZD4 through the resistor R2. Whereas, the modulating voltage from MPX is applied to another varactor diode pair ZD1 and ZD2 through the resistor R2. ZD1 and ZD2 are then biased to a fixed voltage from the DC amplifier that does not have DC blocking capacitance as Figure 35a. These arrangements provides a fixed operating point on the varactor diodes for the modulating voltage through out the entire control voltage from the synthesizer, hence, a constant capacitance change or say the constant frequency deviation for the same amount of modulation voltage under any conditions can be achieved.

8.3.5 Synthesizer Phase Detector
Built-in phase/frequency detector inside 74HCT4046 is used to implement the phase comparator in this VHF synthesizer. The tri-state configuration of this phase/frequency detector has three different states; they are high (+5V), Low (0V) and open-circuit (High-impedance).

Figure 8.37a and 8.37b show it’s simplified circuit and its inputs and output signals. In Figure 8.37b, when Signal Input phase lags Reference Signal, PFD output goes high starting from the rising edge of the Reference signal to the rising edge of Signal Input. When Signal Input phase leads Reference Signal, PFD output goes low starting from the rising edge of the Reference signal to the rising edge of Signal Input. When both inputs have the same phase, both CMOS transistor Q1 & Q2 are off and the output becomes high impedance. In this way, the output is proportional to the phase difference and the output characteristic is shown in Figure 8.37c. Hence, the gain of this type of phase detector (Kp) is (VOH – VOL)/4 or 5V/4 = 0.398V/Rad for TTL compatible logic gates.


8.3.6 Synthesizer Loop Filter
The main task of a loop filter is used to smooth the output pulses of the PFD and provides the dc control voltage to control the frequency of the VCO. Equation 5-6 shows its closed-loop transfer function and is very important to the characteristic of the PLL response. 2nd order active filter as shown in Figure 8.38 is used as the loop filter.

Using the Laplace transform and the transfer function KF(s) for the active filter in Figure 8.38 is: KF(s) = { 1 + sC1 x R2 } / {sC1(R1 + R2)} = {1 + s2} / s1 (8-5) Where: 1 = C1(R1 + R2) and 2 = C1 x R2 (8-6) When Equation (8-1) is substituted into the PLL closed-loop transfer function, Equation (5-6). R1 & R2 can be solved and two equations are derived as: R1 = K / (n2 x N x C1) (8-7) R2 = 2 / (n x C1) (8-8) Where K = Phase Detector gain x VCO gain = Kp x Kv, n = Natural angular frequency of the loop, and  = Damping Factor.

The following Table 8.5 tabulates the 2nd order PLL parameters using a freeware program from KD9JQ. Table 8.5 2nd order PLL filter parameters Parameters Values Reasoning Bandwidth 5Hz ¼ of lowest modulating frequency Damping Factor () 0.7071 Max 20% overshoot VCO Gain (Kv) 2.5MHz/V (54MHz – 44MHz) / 4V Phase Detector Gain (Kp) 0.398 V/Rad VDD/4 (refer Fig 37c) Reference Frequency (Fr) 12.5KHz Division Ratio (N) 4000 50MHz/12.5KHz (Midband) R1 146K ohm 1 = (R1 + R2) C1 = 6.956s 2 = C1 x R2 = 94ms C1 47uf R2 2K Ohm Natural Angular Frequency (n) 4.88 Reference Frequency Suppression -71.267dB VCO Noise Suppression -27.539dB In this case of frequency modulation, the addition of a modulating signal is after the loop filter acts to directly modulate the VCO frequency provided that the loop response is sufficiently slow to prevent the loop filter output from opposing the modulating signal. In other words, the loop maintains a constant average frequency whilst allowing rapid modulation around the center carrier frequency. This gives rise to a high-pass characteristic with a peak frequency deviation of fc = Vfm Kv at high modulation frequencies.

In practical situation, the bandwidth of the loop is chosen to be around 1/4 to 1/5 of the lowest modulating frequency. Hence, loop bandwidth is therefore (20Hz x ¼) 5Hz. However, R2 in the loop filter allows a little ripple to pass and achieves a damping effect to the loop response, which stops the whole loop from probable oscillation. In practical situation, to compromise sufficient response to the step or transient input between the loop stability. The optimum damping factor  of 0.707 is quite a good choice for FM modulators, which gives rise to a Butterworth high-pass response producing the maximally flat characteristic as in Figure 39b. The loop filter response and the FM characteristic of a 2nd order PLL are illustrated in Figure 8.39a and 8.39b respectively.

8.4 RF AMPLIFIER
The RF output power from the VCO is around 50mw, a range of up to 500 metres in free space is possible and which is more than the actual needs of this project for low power in-house transmission.
However, for investigation purpose, further development and enhancing the coverage range, I decided to boost this RF power up to 1W. Hence, an additional RF power amplifier is designed to meet these special requirements. The block diagram of this RF amplifier consists of three main parts as shown in Figure 8.40; they are RF Driver Stage, RF Power Stage and finally the Harmonics Filter.

8.4.1 RF Driver Stage
The completed circuit diagram of the RF amplifier is as shown in the below Figure 8.41.

Transistor Q6 is biased and operated in class-A mode, which amplifies the signal coming from the VCO. It features regeneration to provide high gain and enhances 88MHz to 108MHz pass-band shaping. The regeneration is derived from the Q6’s emitter feedback resistor R30 and the decoupling capacitor C40, the time constant R30 x C40 provides a high gain beyond the frequency of 32MHz and lower frequency filtering. The gain vs frequency graph can be illustrated in Figure 8.42.











8.4.2 RF Power Stage
The final stage 1W VHF class-C amplifier design is referring to the typical tuned power amplifier as shown in Figure 8.43a, which can be obtained from most manufacturers’ data sheet. However, this high gain amplifier is rather unstable in the practical environment due to excessive components lead length, inter-stage capacitive and inductive coupling, which is easily prone to the lower frequency oscillation around 1MHz. Figure 8.43b shows the typical characteristic of some VHF transistors’ gain vs frequency. The amplifier has a power gain of 10dB at 100MHz and has a gain of more than 30dB at 1MHz. With this high gain in the lower frequency region, the amplifier will easily be oscillating parasitically. Since the final stage amplifier is operated in Class-C, which is very non-linear and acts as a mixing device. Finally, the 1MHz self-oscillating frequency will amplitude-modulate the working carrier frequency and gives out a wide spectrum of spurious VHF frequencies, which interferes the near-by stations.

Typical VHF transistor’s gain Vs frequency Figure 8.44 demonstrates the frequency spectrum of an unstable VHF amplifier working in 100MHz carrier frequency.

To stop this unwanted oscillation and interference of the Class-C VHF amplifier, some damping resistors are inserted into the input and output terminals of this power transistor to reduce the lower frequency gain as shown in Figure 8.45.

In this exchange of stability, overall power gain of the VHF amplifier will be lowered.





















8.4.3 Harmonics Filter

Since the final stage of the RF amplifier is operated in Class-C mode, which is very non-linear and the output is rich in harmonics and will cause severe interference. For example, when the 100MHz carrier frequency is being transmitted, there will be some spurious signals 300MHz, 500MHz … etc representing the odd-harmonics of the carrier signal contaminating the spectrum. This means that a high attenuation harmonic low pass filter (LPF) must be required to reduce the harmonics to an acceptable level before being passed to the antenna. Hence the characteristic of the harmonic filters should have low attenuation to the pass band but a very steep and high attenuation to the unwanted harmonics. In order to reduce the absolute power of the harmonic radiation, the acceptable suppression of the third harmonic is about –60dbc. Recall the Table 8.3 that I mentioned in Section 8.2.1 (15KHz LPF design), Chebyshev LPF is the best choice for this situation as the phase and amplitude ripples within the pass-band was not critical, however it can be optimized in design, and the Chebyshev gives better and steep stop-band attenuation than a Butterworth. Normally, higher order filters give a better results in harmonics suppression, whereas in compromising the additional insertion loss and the acceptable harmonic suppression.

Figure 8.46 shows how filter frequency response graphically. In this case, a low pass filter indicates the allowable range of filter gain (i.e. the ripple) in the pass-band, the minimum frequency at which the response leaves the pass-band, the maximum frequency at which the response enters the stop-band, and the minimum attenuation in the stop-band.

The cut-off frequency of this LPF is design to be 113MHz, which gives a 5MHz margin to the highest 108MHz operating frequency. In single frequency operation, greater pass-band ripples can be acceptable for exchanging a sharpness of knee with rapid stop-band attenuation. But in this case, the operating frequency is starting from 88MHz up to 108MHz; 0.1dB - 0.2dB pass-band ripple is optimized to have an acceptable steepness in the transition region and minimum loss in the pass-band.

The LPF in this design is implemented by a 7-pole Chebyshev low pass filter, which is constructed by 7 reactive elements; containing 3 inductors and 4 capacitors and both the input and output impedance is designed to be 50 ohm and the arrangement is shown in Figure 8.47.


Using the Faisyn shareware filter design utility (Figure 8.48) downloaded from FaiSyn RF Design Software Home Page allows these trade-offs to be easily investigated and simulated.

The 113MHz cutoff frequency with 0.2dB optimized 7-pole Chebyshev low pass filter’s response curve and phase characteristic are shown in Figure 8.49a and 8.49b respectively.

The major difficulty is to obtain the required value of inductors for the filter. Fortunately, the required inductance can be calculated and wound using the following formula, which is obtained from Glade Wilcox' 1960 classic, "Basic Electronics." L = (0.2 x A2 x N2) / (3A + 9B + 10C) (8-9) Where: L= Inductance in micro-henrys, N= Number of turns, A= Diameter of the coil (inches), B = length of the winding (inches) & C = Pitch of winding (inches) Another useful shareware COIL V1.7 downloaded from Saratoga Software Corporation facilitates the calculation progress and the following Figure 8.38 shows the snapshot of the program and the value of coils using internal diameter=6mm, pitch= 1.8mm with SWG#18 copper wire.

From Figure 8.50, 5 turns of winding gives a inductance 105.7nH, which is very close to the calculated values of 100nH & 115nH in Faisyn and will be used in this project.

The coil can be constructed by winding the copper wire on a 6mm drill bit and then extends the coil length to 9mm. However, the standard value capacitors available in the market replaces the odd calculated value and the practical components of the LPF are tabulated on Table 8.6.

The practical schematic diagram for the 113MHz 7-element Chebyshev LPF is shown in Figure 8.51.

There have some value adjustment in the practical and calculated components value, and this may lead to filter performance changes. However, stand-alone filter alignment will not be performed individually, instead a completed RF amplifier performance and functional check will be performed to confirm the harmonic suppression characteristic as a whole.

These inductors are mounted on the PCB and mutually orientated at 900 to each other as shown in Photo 8.5. This is an attempt to reduce the mutual inductive coupling between the inductors, this tending to degrade the stop-band attenuation.






8.5 ANTENNA
The antenna used in this RF Audio Link Base Unit is a ¼ λ vertical orientated whip antenna, which is constructed by soldering a Φ3mm with 75mm long brass rod on a UHF type PL259 socket center pin and isolated from the ground sleeve as shown in Figure 8.52.

This type if antenna is commonly used at VHF and UHF. The length of this antenna is determined by: Length (L) = (Speed of Light / Radiation Frequency) x ¼ (8-9) Where Radiation Frequency = (88MHz x 108Mhz)1/2 = 97.5MHz Approx: = 100MHz = (300000000m / 100MHz) x ¼ = 75cm The ¼ λ antenna is connected to the RF Amplifier’s output via a short RG58U feeder and a UHF SO239 socket as shown in Photo 8.6 and Photo 8.7.

The center core of the RG58U feeder is connected to the vertical antenna conductor, while its screen is connected to the chassis ground. This connection means that the antenna is fed at a current maximum as shown in Figure 8.53.

The conductive ground plane acts as a reflector, which makes the antenna’s characteristic as a normal dipole antenna, which has a low angle of radiation and little power is wasted by radiation in the upward direction. In view of the radiation pattern as shown in Figure 8.54a and 8.54b, this antenna is very suitable for our application.




























8.5 POWER SUPPLY
This RF Link System involves mixed design of audio, RF and digital signal processing circuitries; a poorly designed power supply can result in poor performance and unintended interference. In many literatures and electronic devices application notes, they usually claim that there is no real advantages gained by using separate analog and digital supplies. However, in-order to ease the power sourcing, facilitate the clustering of different modules within the RF Audio Link Base Unit and further reduce the interference among modules, separate analog and digital regulated power supplies will be employed. Linear supplies were selected instead of a switching supply to prevent switching noise from being a problem. It is more important that these supplies be as clean as possible to reduce coupling of supply noise to the output.

In my design, individual and separated pre-regulated power supplies for analogue/audio circuits and digital/RF circuits are configured as the following Figure 8.55a and housed in a separate aluminium cabinet. The power supply cabinet and the RF Link Base Unit cabinet are interconnected by multi-core detachable power cable (Figure 8.55b) and the actual connection is shown as Photo 8.8

Figure 8.56 is the complete circuit diagram of the Power Supply. The +/- 12V regulated supply for MPX and Remote Control RX Unit is shown in Photo 8.9.

To further reduce the inter-coupling through the power bus that sourcing the different modules and improve the power purity and supply rejection ratio, individual regulators are introduced and designed for these purposes. Proper power supply decoupling mechanism, such as ∏ (C-L-C and C-R-C) filters will be employed at each supply pin in different modules to maximize power supply rejection. The modules that require additional ON-PCB regulators are tabulated on Table 8.7 and details refer to schematic diagrams in Appendix (I).




































CHAPTER 9: PCB DESIGN


The printed circuit boards are laid out from the schematics by hand using the ProTel ESAYTRAX software. All PCB are designed and fabricated leaving as much area on the component side and bottom side as possible for a contiguous ground plane. The process generally involves creating components, placing them, creating connections, and then routing the connections. Automatic routing is useful in some areas but in general the results are not very good. Manual routing is used to produce most boards. Surface-Mount parts are ideal for compact PCB design and high frequency circuits but are very difficult to fabricate without any dedicated tools and instrument in home-brew process. The final decision is to use general top-mount parts and single-sided PCB / double-sided PCB, and reserves a large grounding area spreading around the PCB for achieving optimal performance and low impedance grounding. The analog and digital grounds on the PCB modules of MPX, 19KHz/38KHz Signal Generator, and VHF Synthesizer and RF Amplfier, etc have been separated to optimize the management of the return currents in the system, see examples of PCB design in Figure 9.1b and Figure 9.2b.

A HEAD-TO-TAIL PCB design strategy is adopted, which means that the design follows the logical and the physical signal flow paths and never allowing input part and output part closed together, and each functional parts within the PCB are clustered so that interference among clusters and components, and unwanted feedback and current loop can be eliminated, see examples of PCB components layer design in Figure 9.1a and Figure 9.2a. The results also confirm the success of these PCB design strategy and no unwanted oscillations are observed. The Schematic and PCB design for each module is shown in the Appendix (II).















CHAPTER 10: MEASUREMENT


This RF Stereo Audio Link is integrated by several important element, the individual characteristics and stability will influence the final outcome and performance of this product as a whole. Hence, before the measurement of the performance of the RF link, measurement for some critical building blocks will be done first. For easy interpretation and understanding of the measurement, visualization by using simple graphs and pictures will be used to aid in confirming the methodology I adopted.

10.1 STEREO SIGNAL ENCODER
Stereo Signal Encoder is the main and core unit for audio signal processing inside this RF link. The characteristic of its constituent parts will directly affect the ultimate performance of this audio link, such as frequency response, S/N, distortion and channel separation. This section describes the setup procedures to optimize the performance of the Stereo Signal Encoder, which includes:  Calibration of 4.096MHz reference crystal used in Pilot & Subcarrier Generator.  Optimizing the RLC resonant filter in Pilot & Subcarrier Generator to generate a stable and clean 19KHz & 38KHz signal.  Measurement of 15KHz LPF characteristic in MPX  Measurement of (L-R) signal in MPX’s signal Matrix  Measurement of channel separation

10.1.1 4.096MHz REFERENCE CRYSTAL CALIBRATION






























10.1.2 PURITY & STABILITY OF 19KHz & 38KHz SIGNAL










































10.1.3 MEASUREMENT OF 15KHz LPF






















10.1.4 MEASUREMENT OF CHARACTERISTIC OF (L-R) SIGNAL & ON-BOARD CHANNEL SEPARATION.




























According to the above measurement, the on board channel separation can be approximated as: On Board Channel Separation = V2 / V1 or 20log(V2/V1) Hence the channel separation before the (L-R) signal modulates the 38KHz sub-carrier; can be illustrated by the following Figure 10.5.





10.3 VHF SYNTHESIZER AND RF AMPLIFIER
The objectives of the following measurement are to measure the tuning range of the synthesized carrier frequency, it’s stability, and the spectral purity as well as the maximum radiated RF power. Moreover, the spectral purity around the center frequency is my major concern, it also reflects how success of this project. If the output from the antenna or say the RF amplifier contains a lot of spurious frequencies spread over the frequency spectrum, a great interference will jam other stations and will get many complaints.

10.3.1 6.4MHz REFERENCE CRYSTAL CALIBRATION













































10.3.1 TUNING RANGE OF THE VHF SYNTHESIZER










































10.3.2 MAXMIUM POWER MEASUREMENT














10.3.3 SPECTRAL PURITY MEASUREMENT


From the measurement, it shows unnoticeable 2nd, 3rd harmonic or even higher order harmonics of the center frequency. The output is very clean down to the measurement noise floor. This measurement surprises me that whether I made a wrong setting or the perfection of the harmonic filter I constructed. However, time is limited and with this RF Amplifier with 0.95W maximum output power, I think that the transmitter is unlikely to cause any problems. This was also confirmed by a real transmission that causing no interference to other FM stations and TVs.



10.4 REAL TRANSMISSION AND FUNCTIONAL TEST

The real transmission and functional test of the RF Stereo Link use the following sets of equipment, a portable FM receiver Model: SRF-M806, the Hand-held Remote Control Unit, a CD player Model: Shinco DVD-868 and the RF Audio Link Base Unit as shown and the connection is shown in Figure 10.8. The tests include control link and audio link range test, control link and audio link functional test.




















10.4.1 CONTROL LINK FUNCTIONAL TEST

The objective of this test is to check if the source select button on the Remote Control TX unit being pressed are correctly decoded by the Remote Control RX unit and activates the correct source input. Since LEDs are installed for the aid of input selection indicator as shown in Figure 10.9a, hence functionality of the control can be easily confirmed. The test is carried out at a distance of 1m between the base unit and the remote control unit and the functional test is confirmed fine (reliability over 99%) by holding the button until the TX Enable LED dims, figure 10.9b. This confirms my design for the debouncing mechanism in section 7.1.3, that the average button pressing duration of 700ms. However, pressing less than 700ms or releases before the Tx Enable LED dims will cause false trigger for the input selection, hence this situation should be avoided.

10.4.2 CONTROL LINK RANGE TEST






10.4.3 AUDIO LINK RANGE TEST

Since the audio link is working in traditional FM broadcasting band, carefully selection of idle or unused channel is very important to avoid unexpected interference to legal ON-AIR programme transmission. In my home area (Sheung Shui), channel at 104.2MHz is idle and suitable for this range testing. In addition, 0.95W of RF output power is quite high in this application, so I limit the maximum transmission duration as short as possible (about 10 mins) to avoid complaints. In this situation, the maximum range is uncertain for this measurement and the result is just for reference.


CHAPTER 11: CONCLUSION AND SUGGESTION

11.1 CONCLUSION
In this project report, some designs and strategies on constructing Stereo Signal Encoder, frequency synthesizer, frequency modulation through PLL method and RF amplifier are examined individually and then constituted in a workable entity, and the main objective is to confirm the improvement in the performance that compared with the existing marketed RF headphones.

A simple but effective RLC parallel resonant filter was designed to provide a clean signal from a raw square wave, which was proved to be successful. Although using PLL method to generate 19KHz and 38KHz is expensive and larger in size, it is undeniable that a very stable and accurate frequency can be generated.

Dual channeled and mirrored design, using high quality and wide-bandwidth-op-amps, and the clustered and grounding method in PCB design for the audio processing circuit (MPX) again proved to be very successful in the enhancement of the channel separation.

Various circuit techniques to minimize spurious harmonics in RF amplifiers, such as minimizing the power gain by adding damping resistors at the input and output of the RF transistor, RF 7-pole Chebyshev harmonic filter design using FAISYN and self-constructed coils were also confirmed to be workable and successful in harmonics and interference elimination by the spectral purity check. Regulated power supplies and sufficient power decoupling for each module were also tested to be useful and crucial in ripple suppression, noise and overall performance of this RF link.

The 103Mhz to 105.7MHz tuning range of the VHF synthesizer was not matched to my project expectation and design; it should cover the entire FM band of 88MHz to 108MHz. However, which is the most difficult part that I experienced in this project. The narrow tuning range of the VHF Synthesizer may be due to the following reasons:

• Lack of sufficient data on the characteristic of the older type VHF varactor diode KV1310, such as capacitance vs control bias. The rough data of KV1310 shown in this report was obtained from my friend and had not been verified.
• Improper design value for the VCO’s resonant circuit; inductance and capacitance of air-coil and capacitor.
• +5V DC supply voltage for the synthesizer’s DC amplifier after the loop filter is too low to provide sufficient control voltage for biasing the entire workable area of the KV1310, i.e. capacitance changes is limited.
• Lack of experience and knowledge in dealing with PLL, especially the design of an optimal loop filter. Further investigation of wide-band synthesizer must be performed.

In reviewing the result of this project, I appreciate myself that I can complete this challenging task within a short period of 8 months, and the final result is good and surprising. Within this 8-months of project work and study, I have learnt many things from the new realm of RF communications that I haven’t experienced before. My pass experience of audio amplifiers design and construction is a major factor that alleviates the problem in familiarizing with the RF circuits. Techniques and knowledge in designing PCB involving audio frequency, radio frequency, digital circuit, mixed mode of analogue and digital, and grounding method in RF are gained. Internet and World-Wide-Web is found to be a immense data-base of information, searching the web intelligently gives me an opportunity to obtain information about new RF technology and other people’s works and experience that I can refer to.

Last but not least, this project course is challenging and a wonderful experience for me. I subjectively think that I have done a good job but need further investigation and study for the subject of RF communications.


11.2 SUGGESTION FOR FURTHER DEVELOPMENT ON THE RF AUDIO LINK
In this project, the cost and size of the final product was not my main concern and I mostly concentrated in seeing how my designs and methods could be improving. Frankly speaking, this product is not marketable for its size and cost. According to the market trend, it should be compact in size and cheap. In considering this issue and compromising with the channel separation, a simple design in term of component count should be adopted.

Analogue design of the stereo signal encoder is a straight-forward design, which has the advantage of higher channel separation that compared with the existing low-end MPX chip-sets available in the market but more complex circuit design, expensive, larger size and difficult alignment procedures. Hence, focusing in these problems of analogue design, digital method in stereo signal encoding is proposed to enhance the performance of the channel separation, simplify and cut-down the cost of the final product. The suggested configuration of the digital MPX unit is shown as Figure 10.10.


In the suggested Digital MPX, all the constituent parts of an analogue MPX is embedded in a DSP and the stereo encoding is performed digitally within the DSP. An analogue15KHz anti-aliasing filter is added before the DSP for the analogue audio input and a 53KHz LPF is added after the DAC for remove harmonics before entering the composite signal to the exciter. The entire building blocks of the transmitter may look like as Figure 10.11.










REFERENCES

1. Arthur B. Williams & Fred J. Taylor, Electronic Filter Design Handbook, Third Edition, Mcgraw Hill, 1995.

2. Green / Bourque, Trouble Shooting, Servicing, And Theory of AM, FM, & FM Stereo Receivers, 2nd Edition. Prentice Hall 1980.

3. D C Green, Radio Systems for Technicians. Second Edition. Longman 1995.

4. Gary M. Miller, Modern Electronic Communication. Fifth Edition. Prentice Hall, 1996.

5. George Kennedy, Electronic Communication Systems. Second Edition. Mcgraw Hill,1981.

6. Henrik Gutsch, Wireless Fits Keyless Entry Applications. Wireless Systems Design, January 2000.

7. Ian Poole, Basic Radio Principles & Technology. Newnes, 1998.

8. Joseph J. Carr, Secrets of RF Circuit Design. Second Edition. Mcgraw Hill, 1997

9. Joseph J.Carr, Practical Antenna Handbook. Second Edition. McGraw Hill, 1994.

10. Motorola Communications Device Data, REV 3. Motorola 1993.

11. Motorola High-Speed CMOS Data, REV 5. Motorola 1993.

12. Paul Horowitz & Winfield Hill, The Art of Electronics. Second Edition. Cambridge University Press, 1996.

13. P. V. Brennan, Phase-Locked Loops: Principles and Practice. Macmillan, 1996.

14. Steve Winder, Filter Design. Newnes, 1997.

15. Tobey, Graeme & Huelsman, Operational Amplifiers Design and Applications. Mcgraw Hill, 1986

16. Ulrich L. Rohde & T. T. N. Bucher, Communications Receivers Principle & Design. McGraw Hill, 1988.

17. William I. Orr, W6SAI, Radio Handbook. Twenty-third Edition. SAMS, 1991.

18. William Schweber, Electronic Communication Systems. Third Edition. Prentice Hall. 1999.

19. 電子電路設設計盲點突破經典, 陳連春編譯 1994.

4 comments:

  1. Useful information shared. I am very happy to read this article..Thanks for giving us nice info. Fantastic walk-through. I appreciate this post. There is lot of articles on the web about this. But I like yours more, although i found one that’s more descriptive. cell phone detector

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  2. This is my final year project . I just want to document what I have done. thanks.

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